Texas Instruments 的 LM2876 规格书

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LM2876

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LM2876 Overture™ Audio Power Amplifier Series

High-Performance 40W Audio Power Amplifier w/Mute

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1FEATURES DESCRIPTION

The LM2876 is a high-performance audio power

23• 40W Continuous Average Output Power Into amplifier capable of delivering 40W of continuous

8Ωaverage power to an 8Ωload with 0.1% THD+N from

• 75W Instantaneous Peak Output Power 20Hz–20kHz.

Capability The performance of the LM2876, utilizing its Self

• Signal-to-Noise Ratio ≥95 dB(min) Peak Instantaneous Temperature (°Ke) (SPiKe)

• An Input Mute Function protection circuitry, puts it in a class above discrete

and hybrid amplifiers by providing an inherently,

• Output Protection From A Short to Ground Or dynamically protected Safe Operating Area (SOA).

to the Supplies Via Internal Current Limiting SPiKe protection means that these parts are

Circuitry completely safeguarded at the output against

• Output Over-Voltage Protection Against overvoltage, undervoltage, overloads, including shorts

Transients From Inductive Loads to the supplies, thermal runaway, and instantaneous

temperature peaks.

• Supply Under-Voltage Protection, Not Allowing

Internal Biasing to Occur When |VEE| + |VCC|≤The LM2876 maintains an excellent signal-to-noise

12V, Thus Eliminating Turn-On and Turn-Off ratio of greater than 95dB (min) with a typical low

Transients noise floor of 2.0μV. It exhibits extremely low THD+N

values of 0.06% at the rated output into the rated

• 11-Lead PFM Package load over the audio spectrum, and provides excellent

• Wide Supply Range 20V - 72V linearity with an IMD (SMPTE) typical rating of

0.004%.

APPLICATIONS

• Component Stereo

• Compact Stereo

• Self-Powered Speakers

• Surround-Sound Amplifiers

• High-End Stereo TVs

Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

2Overture is a trademark of Texas Instruments.

3All other trademarks are the property of their respective owners.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

l IEXAS INSTRUMENTS V+ TNPUT o— RTN 4—W—m ‘0 “1 OUTPUT l1 IO NC WN+ Vm’ MUTE GND NC NCO‘) v- OUTPUT NC v+ O LM2876 ANmbUImum

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Typical Application

* Optional components dependent upon specific design requirements. Refer to the External Components Description

section for a component functional description.

Figure 1. Typical Audio Amplifier Application Circuit

Connection Diagram

†Connect Pin 5 to V+for Compatibility with LM3886.

*Preliminary: Call your local Texas Instruments' sales rep. or distributor for availability.

Figure 2. Top View

See Package Number NDJ0011B

for Staggered Lead Non-Isolated Package

or NDA0011B for Staggered Lead Isolated Package

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These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

ABSOLUTE MAXIMUM RATINGS (1) (2)(3)

Supply Voltage |V+|+|V−| (No Signal) 72V

Supply Voltage |V+|+|V−| (Input Signal) 70V

Common Mode Input Voltage (V+or V−) and |V+| + |V−|≤60V

Differential Input Voltage 60V

Output Current Internally Limited

Power Dissipation (4) 125W

ESD Susceptibility (5) 3000V

Junction Temperature (6) 150°C

Soldering Information NDJ Package (10 seconds) 260°C

Storage Temperature −40°C to +150°C

θJC 1°C/W

Thermal Resistance(7)

θJA 43°C/W

(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for

which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical

specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the

Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication

of device performance.

(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and

specifications.

(3) All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified.

(4) For operating at case temperatures above 25°C, the device must be derated based on a 150°C maximum junction temperature and a

thermal resistance of θJC = 1.0 °C/W (junction to case). Refer to the Thermal Resistance figure in the APPLICATION INFORMATION

section under THERMAL CONSIDERATIONS.

(5) Human body model, 100 pF discharged through a 1.5 kΩresistor.

(6) The operating junction temperature maximum is 150°C, however, the instantaneous Safe Operating Area temperature is 250°C.

(7) The LM2876T package NDJ0011B is a non-isolated package, setting the tab of the device and the heat sink at V−potential when the

LM2876 is directly mounted to the heat sink using only thermal compound. If a mica washer is used in addition to thermal compound,

θCS (case to sink) is increased, but the heat sink will be isolated from V−.

OPERATING RATINGS (1) (2)(3)

Temperature Range

TMIN ≤TA≤TMAX −20°C ≤TA≤+85°C

Supply Voltage |V+| + |V−| 20V to 60V

(1) All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified.

(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for

which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical

specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the

Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication

of device performance.

(3) Operation is specified up to 60V, however, distortion may be introduced from SPiKe Protection Circuitry if proper thermal considerations

are not taken into account. Refer to the THERMAL CONSIDERATIONS section for more information. (See SPiKe Protection Response.)

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ELECTRICAL CHARACTERISTICS (1) (2)

The following specifications apply for V+= +30V, V−=−30V, IMUTE =−0.5 mA with RL= 8Ωunless otherwise specified. Limits

apply for TA= 25°C.

Symbol Parameter Conditions LM2876 Units

(Limits)

Typical(3 Limit(4)

)

20 V (min)

|V+| + |V−| Power Supply Voltage (5) Vpin7 −V−≥9V 18 60 V (max)

Pin 8 Open or at 0V, Mute: On Current out of Pin 115 80 dB (min)

AMMute Attenuation 8 > 0.5 mA, Mute: Off

PO(6) Output Power (Continuous Average) THD + N = 0.1% (max) f = 1 kHz; f = 20 kHz 40 25 W (min)

Peak POInstantaneous Peak Output Power 75 W

THD + N Total Harmonic Distortion Plus Noise 25W, 20 Hz ≤f≤20 kHz AV= 26 dB 0.06 %

SR (6) Slew Rate (7) VIN = 1.2 Vrms, f = 10 kHz, Square-Wave, RL= V/μs (min)

9 5

2 kΩ

I+(8) Total Quiescent Power Supply Current VCM = 0V, Vo= 0V, Io= 0A 24 50 mA (max)

VOS(8) Input Offset Voltage VCM = 0V, Io= 0 mA 1 10 mV (max)

IBInput Bias Current VCM = 0V, Io= 0 mA 0.2 1 μA (max)

IOS Input Offset Current VCM = 0V, Io= 0 mA 0.01 0.2 μA (max)

IoOutput Current Limit |V+| = |V−| = 10V, tON = 10 ms, VO= 0V 4 3 A (min)

|V+–VO|, V+= 20V, Io= +100 mA 1.5 4 V (max)

Vod(8) Output Dropout Voltage (9) |VO–V−|, V−=−20V, Io=−100 mA 2.5 4 V (max)

V+= 30V to 10V, V−=−30V, VCM = 0V, Io= 0 125 85 dB (min)

PSRR (8) Power Supply Rejection Ratio V+= 30V, V−=−30V to −10V, VCM = 0V, Io= 0 110 85 dB (min)

V+= 50V to 10V, V−=−10V to −50V, VCM = 20V

CMRR (8) Common Mode Rejection Ratio 110 75 dB (min)

to −20V, Io= 0 mA

AVOL(8) Open Loop Voltage Gain |V+| = |V−| = 30V, RL= 2 kΩ,ΔVO= 40V 115 80 dB (min)

GBWP Gain-Bandwidth Product |V+| = |V−| = 30V, fO= 100 kHz, VIN = 50 mVrms 8 2 MHz (min)

IHF—A Weighting Filter, RIN = 600Ω(Input

eIN (6) Input Noise 2.0 8 μV (max)

Referred)

PO= 1W, A-Weighted, Measured at 1 kHz, RS=98 dB

25Ω

PO= 25W, A-Weighted, Measured at 1 kHz, RS

SNR Signal-to-Noise Ratio 112 dB

= 25Ω

Ppk= 75W, A-Weighted, Measured at 1 kHz, RS117 dB

= 25Ω

60 Hz, 7 kHz, 4:1 (SMPTE) 0.004

IMD Intermodulation Distortion Test %

60 Hz, 7 kHz, 1:1 (SMPTE) 0.006

(1) All voltages are measured with respect to the GND pin (pin 7), unless otherwise specified.

(2) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for

which the device is functional, but do not ensure specific performance limits. Electrical Characteristics state DC and AC electrical

specifications under particular test conditions which ensure specific performance limits. This assumes that the device is within the

Operating Ratings. Specifications are not ensured for parameters where no limit is given, however, the typical value is a good indication

of device performance.

(3) Typicals are measured at 25°C and represent the parametric norm.

(4) Limits are specified to Texas Instruments' AOQL (Average Outgoing Quality Level).

(5) V−must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled.

(6) AC Electrical Test; refer to Test Circuit #2 .

(7) The feedback compensation network limits the bandwidth of the closed-loop response and so the slew rate will be reduced due to the

high frequency roll-off. Without feedback compensation, the slew rate is typically larger.

(8) DC Electrical Test; refer to Test Circuit #1 .

(9) The output dropout voltage is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs Supply Voltage graph in the

TYPICAL PERFORMANCE CHARACTERISTICS section.

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l TEXAS INSTRUMENTS wan/m mom 0 DU'PUY man 4950 m 1m Ru 2““ c, SUpF Ru 20m SDJRCE 70.5 ma

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Test Circuit #1

(DC Electrical Test Circuit)

Figure 3.

Test Circuit #2

(AC Electrical Test Circuit)

Figure 4.

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SINGLE SUPPLY APPLICATION CIRCUIT

*Optional components dependent upon specific design requirements. Refer to the External Components Description

section for a component functional description.

Figure 5. Typical Single Supply Audio Amplifier Application Circuit

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l TEXAS INSTRUMENTS v+ O Ik MUTE MD; 045 ouvpuv AN 0— 7w

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Equivalent Schematic

(excluding active protection circuitry)

Figure 6.

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External Components Description(10)

Components Functional Description

1. RIN Acts as a volume control by setting the voltage level allowed to the amplifier's input terminals.

2. RAProvides DC voltage biasing for the single supply operation and bias current for the positive input terminal.

3. CAProvides bias filtering.

4. C Provides AC coupling at the input and output of the amplifier for single supply operation.

5. RBPrevents currents from entering the amplifier's non-inverting input which may be passed through to the load upon

power-down of the system due to the low input impedance of the circuitry when the under-voltage circuitry is off. This

phenomenon occurs when the supply voltages are below 1.5V.

6. CC(1) Reduces the gain (bandwidth of the amplifier) at high frequencies to avoid quasi-saturation oscillations of the output

transistor. The capacitor also suppresses external electromagnetic switching noise created from fluorescent lamps.

7. Ri Inverting input resistance to provide AC Gain in conjunction with Rf1.

8. Ci (1) Feedback capacitor. Ensures unity gain at DC. Also a low frequency pole (highpass roll-off) at: fc= 1/(2πRi Ci)

9. Rf1 Feedback resistance to provide AC Gain in conjunction with Ri.

10. Rf2(1) At higher frequencies feedback resistance works with Cfto provide lower AC Gain in conjunction with Rf1 and Ri. A high

frequency pole (lowpass roll-off) exists at: fc= [Rf1 Rf2 (s + 1/Rf2Cf)]/[(Rf1 + Rf2)(s + 1/Cf(Rf1 + Rf2))]

11. Cf(1) Compensation capacitor that works with Rf1 and Rf2 to reduce the AC Gain at higher frequencies.

12. RMMute resistance set up to allow 0.5 mA to be drawn from pin 8 to turn the muting function off. →RMis calculated

using: RM≤(|VEE|−2.6V)/I8 where I8 ≥0.5 mA. Refer to the Mute Attenuation vs Mute Current curves in the TYPICAL

PERFORMANCE CHARACTERISTICS section.

13. CMMute capacitance set up to create a large time constant for turn-on and turn-off muting.

14. RSN(1) Works with CSN to stabilize the output stage by creating a pole that eliminates high frequency oscillations.

15. CSN(1) Works with RSN to stabilize the output stage by creating a pole that eliminates high frequency oscillations. fc=

1/(2πRSNCSN)

16. L (1) Provides high impedance at high frequecies so that R may decouple a highly capacitive load and reduce the Q of the

series resonant circuit due to capacitive load. Also provides a low impedance at low frequencies to short out R and pass

17. R (1) audio signals to the load.

18. CSProvides power supply filtering and bypassing.

19. S1 Mute switch that mutes the music going into the amplifier when opened.

(10) (Figure 1 and Figure 5)

(1) Optional components dependent upon specific design requirements. Refer to the APPLICATION INFORMATION section for more

information.

OPTIONAL EXTERNAL COMPONENT INTERACTION

Although the optional external components have specific desired functions that are designed to reduce the

bandwidth and eliminate unwanted high frequency oscillations they may cause certain undesirable effects when

they interact. Interaction may occur for components whose reactances are in close proximity to one another. One

example would be the coupling capacitor, CC, and the compensation capacitor, Cf. These two components act as

low impedances to certain frequencies which will couple signals from the input to the output. Please take careful

note of basic amplifier component functionality when designing in these components.

The optional external components shown in Figure 5 and described above are applicable in both single and split

voltage supply configurations.

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l TEXAS INSTRUMENTS I/AT 50.001011 3 E n & 5 3 s o a E 3 B 8 vi = zsuoc 10—220 IC = 25°C o 20 40 so 50 coLLEcIoR-EWTER VOLTAGE (v) TWE(ms) A E E g 5 g 5 ﬂ 3 a 3 E 5i i :' 2 V“ E E 0 I0 20 so 40 50 o 20 40 so so SUFFLV VOLTAGE (xv) CDLLECIOR-[MITTER VOLTAGE (v) so 3 E E 40 m v : : g c: 2 so 5 70v : 9 60V g E 50v L’ 20 4 40V E vS : izuv Vs : ,m g 30v ; 5 V' HJ E o 0.‘ t ‘0 IDO ,50*‘D,30*10,m a m I“ so ‘0 so PULSE WIDTH (ms) OUTPUT vomsz (v)

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TYPICAL PERFORMANCE CHARACTERISTICS

SPiKe

Safe Area Protection Response

Figure 7. Figure 8.

Supply Current vs Pulse Thermal

Supply Voltage Resistance

Figure 9. Figure 10.

Pulse Thermal Supply Current vs

Resistance Output Voltage

Figure 11. Figure 12.

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l TEXAS INSTRUMENTS SUPPLY CURRENT (mA) POWER D‘SS‘PATION (W) \NPUT ms CURRENT (m) an 60 4U 20 20 40 so CDLLECIOFEMITTER VOLTAGE (v) v: = ‘zsnc BO u 50 mo CASE TEMPERATURE (°C) ‘50 >50 0 50 mo TEMPERATURE (0c) 150 OUTPUT (v) PULSE POWER DISS‘PAT‘ON (w) CUPPING VOLTAGE (V) 200 180 150 no 120 I00 an 50 40 20 0.1 t m we PULSE WIDTH (ms) I/AT zovuo kHz m om TWEULS) 5 - — - chppmg Vul‘age(-VEE) , ‘ —- Chppmg Vanagehvcc) / / RL = an X 3 2 IO ‘5 20 25 so SUPPLV VOLTAGE (1v)

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TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Pulse Power Limit Pulse Power Limit

Figure 13. Figure 14.

Supply Current vs

Case Temperature Pulse Response

Figure 15. Figure 16.

Input Bias Current vs Clipping Voltage vs

Case Temperature Supply Voltage

Figure 17. Figure 18.

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TYPICAL PERFORMANCE CHARACTERISTICS (continued)

THD + N

vs THD + N vs

Frequency Output Power

Figure 19. Figure 20.

THD + N vs

Output Power THD + N Distribution

Figure 21. Figure 22.

Output Power vs

THD + N Distribution Load Resistance

Figure 23. Figure 24.

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TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Max Heatsink Thermal Resistance (°C/W)

at the Specified Ambient Temperature (°C)

Maximum Power Dissipation

Supply Voltage

Note: The maximum heat sink thermal resistance values, øSA, in the table above were calculated using a øCS = 0.2°C/W due to thermal

compound. Figure 25.

Power Dissipation vs Power Dissipation vs

Output Power Output Power

Figure 26. Figure 27.

Output Power vs

Supply Voltage IMD 60 Hz, 4:1

Figure 28. Figure 29.

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l TEXAS INSTRUMENTS ‘2000 50 08462 752.04 7.06U00k AP 0.00040 73.303 ‘0 710.45 720.52 ‘MD 60 Hz 000 50 Hz, 7002, m A 70Hz m > 75077 Q m 3 40.92 I E 75900 E 0.1 3 —59.23 L!) 77930 49.54 ”'0‘" 799.7 4000 0,001 7120.0 0.0005 6.000 6.400 6.000 7,200 7500 0000 DJ 1 ‘0 ‘00 5.200 6.600 7.000 7.000 7.000 FREQUENCHM OUTPUT POWER (W) 1 20.000 002101 2000000 “F 00 750.70 7050000 AF 04 an 000 50 Hz 40'0” ' 7 kHz m 3“ E 740.00 : 0.00 :’ E ; —50.00 I l w 4 _ n I 000‘ 0000 WW 4000 0.0001 420.0 20 m0 200 6.000 5.000 6.800 7.200 7.600 0,000 5.200 5.500 7.000 7.400 7.000 FREQUENCV (Hz) FREDUENCHHI) 50 0 m 0.0050: 20 040 50m, 70H:, m 40 o: Mn, 20 kHz E 3 E 0.1 E m s U, 2 an 0,0m 100 v0 : WRMS 0'00. AV : 2500 0.0005 ‘2“ 00m 0.01 00 1 10 Lu t m we OUTPUT POWER (W) PIN 8 MUIE CURRENT (mA)

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TYPICAL PERFORMANCE CHARACTERISTICS (continued)

IMD 60 Hz, 7 kHz, 4:1 IMD 60 Hz, 7 kHz, 4:1

Figure 30. Figure 31.

IMD 60 Hz, 1:1 IMD 60 Hz, 7 kHz 1:1

Figure 32. Figure 33.

Mute Attenuation

IMD 60 Hz, 7 kHz, 1:1 vs Mute Current

Figure 34. Figure 35.

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TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Mute Attenuation

vs Mute Current Large Signal Response

Figure 36. Figure 37.

Power Supply Common-Mode

Rejection Ratio Rejection Ratio

Figure 38. Figure 39.

Open Loop

Frequency Response

Figure 40.

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APPLICATION INFORMATION

GENERAL FEATURES

Mute Function

The muting function of the LM2876 allows the user to mute the music going into the amplifier by drawing less

than 0.5 mA out of pin 8 of the device. This is accomplished as shown in the Typical Application Circuit where

the resistor RMis chosen with reference to your negative supply voltage and is used in conjuction with a switch.

The switch (when opened) cuts off the current flow from pin 8 to V−, thus placing the LM2876 into mute mode.

Refer to the Mute Attenuation vs Mute Current curves in the TYPICAL PERFORMANCE CHARACTERISTICS

section for values of attenuation per current out of pin 8. The resistance RMis calculated by the following

equation:

RM(|VEE|−2.6V)/I8

where

• I8 ≥0.5 mA

Under-Voltage Protection

Upon system power-up the under-voltage protection circuitry allows the power supplies and their corresponding

caps to come up close to their full values before turning on the LM2876 such that no DC output spikes occur.

Upon turn-off, the output of the LM2876 is brought to ground before the power supplies such that no transients

occur at power-down.

Over-Voltage Protection

The LM2876 contains overvoltage protection circuitry that limits the output current to approximately 4Apeak while

also providing voltage clamping, though not through internal clamping diodes. The clamping effect is quite the

same, however, the output transistors are designed to work alternately by sinking large current spikes.

SPiKe Protection

The LM2876 is protected from instantaneous peak-temperature stressing by the power transistor array. The Safe

Operating Area graph in the TYPICAL PERFORMANCE CHARACTERISTICS section shows the area of device

operation where the SPiKe Protection Circuitry is not enabled. The waveform to the right of the SOA graph

exemplifies how the dynamic protection will cause waveform distortion when enabled.

Thermal Protection

The LM2876 has a sophisticated thermal protection scheme to prevent long-term thermal stress to the device.

When the temperature on the die reaches 165°C, the LM2876 shuts down. It starts operating again when the die

temperature drops to about 155°C, but if the temperature again begins to rise, shutdown will occur again at

165°C. Therefore the device is allowed to heat up to a relatively high temperature if the fault condition is

temporary, but a sustained fault will cause the device to cycle in a Schmitt Trigger fashion between the thermal

shutdown temperature limits of 165°C and 155°C. This greatly reduces the stress imposed on the IC by thermal

cycling, which in turn improves its reliability under sustained fault conditions.

Since the die temperature is directly dependent upon the heat sink, the heat sink should be chosen as discussed

in the THERMAL CONSIDERATIONS section, such that thermal shutdown will not be reached during normal

operation. Using the best heat sink possible within the cost and space constraints of the system will improve the

long-term reliability of any power semiconductor device.

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THERMAL CONSIDERATIONS

Heat Sinking

The choice of a heat sink for a high-power audio amplifier is made entirely to keep the die temperature at a level

such that the thermal protection circuitry does not operate under normal circumstances. The heat sink should be

chosen to dissipate the maximum IC power for a given supply voltage and rated load.

With high-power pulses of longer duration than 100 ms, the case temperature will heat up drastically without the

use of a heat sink. Therefore the case temperature, as measured at the center of the package bottom, is entirely

dependent on heat sink design and the mounting of the IC to the heat sink. For the design of a heat sink for your

audio amplifier application refer to the Determining the Correct Heat Sink section.

Since a semiconductor manufacturer has no control over which heat sink is used in a particular amplifier design,

we can only inform the system designer of the parameters and the method needed in the determination of a heat

sink. With this in mind, the system designer must choose his supply voltages, a rated load, a desired output

power level, and know the ambient temperature surrounding the device. These parameters are in addition to

knowing the maximum junction temperature and the thermal resistance of the IC, both of which are provided by

Texas Instruments.

As a benefit to the system designer we have provided Maximum Power Dissipation vs Supply Voltages curves

for various loads in the TYPICAL PERFORMANCE CHARACTERISTICS section, giving an accurate figure for

the maximum thermal resistance required for a particular amplifier design. This data was based on θJC = 1°C/W

and θCS = 0.2°C/W. We also provide a section regarding heat sink determination for any audio amplifier design

where θCS may be a different value. It should be noted that the idea behind dissipating the maximum power

within the IC is to provide the device with a low resistance to convection heat transfer such as a heat sink.

Therefore, it is necessary for the system designer to be conservative in his heat sink calculations. As a rule, the

lower the thermal resistance of the heat sink the higher the amount of power that may be dissipated. This is of

course guided by the cost and size requirements of the system. Convection cooling heat sinks are available

commercially, and their manufacturers should be consulted for ratings.

Proper mounting of the IC is required to minimize the thermal drop between the package and the heat sink. The

heat sink must also have enough metal under the package to conduct heat from the center of the package

bottom to the fins without excessive temperature drop.

A thermal grease such as Wakefield type 120 or Thermalloy Thermacote should be used when mounting the

package to the heat sink. Without this compound, thermal resistance will be no better than 0.5°C/W, and

probably much worse. With the compound, thermal resistance will be 0.2°C/W or less, assuming under 0.005

inch combined flatness runout for the package and heat sink. Proper torquing of the mounting bolts is important

and can be determined from heat sink manufacturer's specification sheets.

Should it be necessary to isolate V−from the heat sink, an insulating washer is required. Hard washers like

beryluum oxide, anodized aluminum and mica require the use of thermal compound on both faces. Two-mil mica

washers are most common, giving about 0.4°C/W interface resistance with the compound.

Silicone-rubber washers are also available. A 0.5°C/W thermal resistance is claimed without thermal compound.

Experience has shown that these rubber washers deteriorate and must be replaced should the IC be

dismounted.

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Determining Maximum Power Dissipation

Power dissipation within the integrated circuit package is a very important parameter requiring a thorough

understanding if optimum power output is to be obtained. An incorrect maximum power dissipation (PD)

calculation may result in inadequate heat sinking, causing thermal shutdown circuitry to operate and limit the

output power.

The following equations can be used to accurately calculate the maximum and average integrated circuit power

dissipation for your amplifier design, given the supply voltage, rated load, and output power. These equations

can be directly applied to the Power Dissipation vs Output Power curves in the TYPICAL PERFORMANCE

CHARACTERISTICS section.

Equation 1 exemplifies the maximum power dissipation of the IC and Equation 2 and Equation 3 exemplify the

average IC power dissipation expressed in different forms.

PDMAX = VCC2/2π2RL

where

• VCC is the total supply voltage (1)

PDAVE = (VOpk/RL)[VCC/π − VOpk/2]

where

• VCC is the total supply voltage and VOpk = VCC/π(2)

PDAVE = VCC VOpk/πRL−VOpk2/2RL

where

• VCC is the total supply voltage (3)

Determining the Correct Heat Sink

Once the maximum IC power dissipation is known for a given supply voltage, rated load, and the desired rated

output power the maximum thermal resistance (in °C/W) of a heat sink can be calculated. This calculation is

made using Equation 4equation (4) and is based on the fact that thermal heat flow parameters are analogous to

electrical current flow properties.

It is also known that typically the thermal resistance, θJC (junction to case), of the LM2876 is 1°C/W and that

using Thermalloy Thermacote thermal compound provides a thermal resistance, θCS (case to heat sink), of about

0.2°C/W as explained in the Heat Sinking section.

Referring to the figure below, it is seen that the thermal resistance from the die (junction) to the outside air

(ambient) is a combination of three thermal resistances, two of which are known, θJC and θCS. Since convection

heat flow (power dissipation) is analogous to current flow, thermal resistance is analogous to electrical

resistance, and temperature drops are analogous to voltage drops, the power dissipation out of the LM2876 is

equal to the following:

PDMAX = (TJmax −TAmb)/θJA

where

•θJA =θJC +θCS +θSA

But since we know PDMAX,θJC, and θSC for the application and we are looking for θSA, we have the following:

θSA = [(TJmax −TAmb)−PDMAX (θJC +θCS)]/PDMAX (4)

Again it must be noted that the value of θSA is dependent upon the system designer's amplifier application and its

corresponding parameters as described previously. If the ambient temperature that the audio amplifier is to be

working under is higher than the normal 25°C, then the thermal resistance for the heat sink, given all other things

are equal, will need to be smaller.

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Equation 1 and Equation 4 are the only equations needed in the determination of the maximum heat sink thermal

resistance. This is of course given that the system designer knows the required supply voltages to drive his rated

load at a particular power output level and the parameters provided by the semiconductor manufacturer. These

parameters are the junction to case thermal resistance, θJC, TJmax = 150°C, and the recommended Thermalloy

Thermacote thermal compound resistance, θCS.

SIGNAL-TO-NOISE RATIO

In the measurement of the signal-to-noise ratio, misinterpretations of the numbers actually measured are

common. One amplifier may sound much quieter than another, but due to improper testing techniques, they

appear equal in measurements. This is often the case when comparing integrated circuit designs to discrete

amplifier designs. Discrete transistor amps often “run out of gain” at high frequencies and therefore have small

bandwidths to noise as indicated below.

Integrated circuits have additional open loop gain allowing additional feedback loop gain in order to lower

harmonic distortion and improve frequency response. It is this additional bandwidth that can lead to erroneous

signal-to-noise measurements if not considered during the measurement process. In the typical example above,

the difference in bandwidth appears small on a log scale but the factor of 10 in bandwidth, (200 kHz to 2 MHz)

can result in a 10 dB theoretical difference in the signal-to-noise ratio (white noise is proportional to the square

root of the bandwidth in a system).

In comparing audio amplifiers it is necessary to measure the magnitude of noise in the audible bandwidth by

using a “weighting” filter (see Note below). A “weighting” filter alters the frequency response in order to

compensate for the average human ear's sensitivity to the frequency spectra. The weighting filters at the same

time provide the bandwidth limiting as discussed in the previous paragraph.

NOTE

CCIR/ARM: A Practical Noise Measurement Method; by Ray Dolby, David Robinson and

Kenneth Gundry, AES Preprint No. 1353 (F-3).

In addition to noise filtering, differing meter types give different noise readings. Meter responses include:

1. RMS reading,

2. average responding,

3. peak reading, and

4. quasi peak reading.

Although theoretical noise analysis is derived using true RMS based calculations, most actual measurements are

taken with ARM (Average Responding Meter) test equipment.

Typical signal-to-noise figures are listed for an A-weighted filter which is commonly used in the measurement of

noise. The shape of all weighting filters is similar, with the peak of the curve usually occurring in the 3 kHz–7 kHz

region as shown below.

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SUPPLY BYPASSING

The LM2876 has excellent power supply rejection and does not require a regulated supply. However, to eliminate

possible oscillations all op amps and power op amps should have their supply leads bypassed with low-

inductance capacitors having short leads and located close to the package terminals. Inadequate power supply

bypassing will manifest itself by a low frequency oscillation known as “motorboating” or by high frequency

instabilities. These instabilities can be eliminated through multiple bypassing utilizing a large tantalum or

electrolytic capacitor (10 μF or larger) which is used to absorb low frequency variations and a small ceramic

capacitor (0.1 μF) to prevent any high frequency feedback through the power supply lines.

If adequate bypassing is not provided the current in the supply leads which is a rectified component of the load

current may be fed back into internal circuitry. This signal causes low distortion at high frequencies requiring that

the supplies be bypassed at the package terminals with an electrolytic capacitor of 470 μF or more.

LEAD INDUCTANCE

Power op amps are sensitive to inductance in the output lead, particularly with heavy capacitive loading.

Feedback to the input should be taken directly from the output terminal, minimizing common inductance with the

load.

Lead inductance can also cause voltage surges on the supplies. With long leads to the power supply, energy is

stored in the lead inductance when the output is shorted. This energy can be dumped back into the supply

bypass capacitors when the short is removed. The magnitude of this transient is reduced by increasing the size

of the bypass capacitor near the IC. With at least a 20 μF local bypass, these voltage surges are important only if

the lead length exceeds a couple feet (> 1 μH lead inductance). Twisting together the supply and ground leads

minimizes the effect.

LAYOUT, GROUND LOOPS AND STABILITY

The LM2876 is designed to be stable when operated at a closed-loop gain of 10 or greater, but as with any other

high-current amplifier, the LM2876 can be made to oscillate under certain conditions. These usually involve

printed circuit board layout or output/input coupling.

When designing a layout, it is important to return the load ground, the output compensation ground, and the low

level (feedback and input) grounds to the circuit board common ground point through separate paths. Otherwise,

large currents flowing along a ground conductor will generate voltages on the conductor which can effectively act

as signals at the input, resulting in high frequency oscillation or excessive distortion. It is advisable to keep the

output compensation components and the 0.1 μF supply decoupling capacitors as close as possible to the

LM2876 to reduce the effects of PCB trace resistance and inductance. For the same reason, the ground return

paths should be as short as possible.

In general, with fast, high-current circuitry, all sorts of problems can arise from improper grounding which again

can be avoided by returning all grounds separately to a common point. Without isolating the ground signals and

returning the grounds to a common point, ground loops may occur.

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“Ground Loop” is the term used to describe situations occurring in ground systems where a difference in potential

exists between two ground points. Ideally a ground is a ground, but unfortunately, in order for this to be true,

ground conductors with zero resistance are necessary. Since real world ground leads possess finite resistance,

currents running through them will cause finite voltage drops to exist. If two ground return lines tie into the same

path at different points there will be a voltage drop between them. The first figure below shows a common ground

example where the positive input ground and the load ground are returned to the supply ground point via the

same wire. The addition of the finite wire resistance, R2, results in a voltage difference between the two points as

shown below.

The load current ILwill be much larger than input bias current II, thus V1will follow the output voltage directly, i.e.

in phase. Therefore the voltage appearing at the non-inverting input is effectively positive feedback and the

circuit may oscillate. If there were only one device to worry about then the values of R1and R2would probably be

small enough to be ignored; however, several devices normally comprise a total system. Any ground return of a

separate device, whose output is in phase, can feedback in a similar manner and cause instabilities. Out of

phase ground loops also are troublesome, causing unexpected gain and phase errors.

The solution to most ground loop problems is to always use a single-point ground system, although this is

sometimes impractical. The third figure below is an example of a single-point ground system.

The single-point ground concept should be applied rigorously to all components and all circuits when possible.

Violations of single-point grounding are most common among printed circuit board designs, since the circuit is

surrounded by large ground areas which invite the temptation to run a device to the closest ground spot. As a

final rule, make all ground returns low resistance and low inductance by using large wire and wide traces.

Occasionally, current in the output leads (which function as antennas) can be coupled through the air to the

amplifier input, resulting in high-frequency oscillation. This normally happens when the source impedance is high

or the input leads are long. The problem can be eliminated by placing a small capacitor, CC, (on the order of 50

pF to 500 pF) across the LM2876 input terminals. Refer to the External Components Description(1) section

relating to component interaction with Cf.

(1) (Figure 1 and Figure 5)

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REACTIVE LOADING

It is hard for most power amplifiers to drive highly capacitive loads very effectively and normally results in

oscillations or ringing on the square wave response. If the output of the LM2876 is connected directly to a

capacitor with no series resistance, the square wave response will exhibit ringing if the capacitance is greater

than about 0.2 μF. If highly capacitive loads are expected due to long speaker cables, a method commonly

employed to protect amplifiers from low impedances at high frequencies is to couple to the load through a 10Ω

resistor in parallel with a 0.7 μH inductor. The inductor-resistor combination as shown in the Typical Application

Circuit isolates the feedback amplifier from the load by providing high output impedance at high frequencies thus

allowing the 10Ωresistor to decouple the capacitive load and reduce the Q of the series resonant circuit. The LR

combination also provides low output impedance at low frequencies thus shorting out the 10Ωresistor and

allowing the amplifier to drive the series RC load (large capacitive load due to long speaker cables) directly.

GENERALIZED AUDIO POWER AMPLIFIER DESIGN

The system designer usually knows some of the following parameters when starting an audio amplifier design:

Desired Power Output Input Level

Input Impedance Load Impedance

Maximum Supply Voltage Bandwidth

The power output and load impedance determine the power supply requirements, however, depending upon the

application some system designers may be limited to certain maximum supply voltages. If the designer does

have a power supply limitation, he should choose a practical load impedance which would allow the amplifier to

provide the desired output power, keeping in mind the current limiting capabilities of the device. In any case, the

output signal swing and current are found from (where POis the average output power):

(5)

(6)

To determine the maximum supply voltage the following parameters must be considered. Add the dropout

voltage (4V for LM2876) to the peak output swing, Vopeak, to get the supply rail value (i.e. ± (Vopeak + Vod) at a

current of Iopeak). The regulation of the supply determines the unloaded voltage, usually about 15% higher.

Supply voltage will also rise 10% during high line conditions. Therefore, the maximum supply voltage is obtained

from the following equation:

Max. supplies

≈± (Vopeak + Vod)(1 + regulation)(1.1) (7)

The input sensitivity and the output power specs determine the minimum required gain as depicted below:

(8)

Normally the gain is set between 20 and 200; for a 40W, 8Ωaudio amplifier this results in a sensitivity of 894 mV

and 89 mV, respectively. Although higher gain amplifiers provide greater output power and dynamic headroom

capabilities, there are certain shortcomings that go along with the so called “gain.” The input referred noise floor

is increased and hence the SNR is worse. With the increase in gain, there is also a reduction of the power

bandwidth which results in a decrease in feedback thus not allowing the amplifier to respond quickly enough to

nonlinearities. This decreased ability to respond to nonlinearities increases the THD + N specification.

The desired input impedance is set by RIN. Very high values can cause board layout problems and DC offsets at

the output. The value for the feedback resistance, Rf1, should be chosen to be a relatively large value (10

kΩ–100 kΩ), and the other feedback resistance, Ri, is calculated using standard op amp configuration gain

equations. Most audio amplifiers are designed from the non-inverting amplifier configuration.

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DESIGN A 25W/8ΩAUDIO AMPLIFIER

Given:

Power Output 25W

Load Impedance 8Ω

Input Level 1V(max)

Input Impedance 100 kΩ

Bandwidth 20 Hz–20 kHz ± 0.25 dB

Equation 5 and Equation 6 give:

25W/8ΩVopeak = 20.0V Iopeak = 2.5A

Therefore the supply required is: ±24.0V @ 2.5A

With 15% regulation and high line the final supply voltage is ±30.36V using Equation 7. At this point it is a good

idea to check the Power Output vs Supply Voltage to ensure that the required output power is obtainable from

the device while maintaining low THD + N. It is also good to check the Power Dissipation vs Supply Voltage to

ensure that the device can handle the internal power dissipation. At the same time designing in a relatively

practical sized heat sink with a low thermal resistance is also important. Refer to TYPICAL PERFORMANCE

CHARACTERISTICS graphs and the THERMAL CONSIDERATIONS section for more information.

The minimum gain from Equation 8 is: AV≥14

We select a gain of 15 (Non-Inverting Amplifier); resulting in a sensitivity of 942.8 mV.

Letting RIN equal 100 kΩgives the required input impedance, however, this would eliminate the “volume control”

unless an additional input impedance was placed in series with the 10 kΩpotentiometer that is depicted in

Figure 1. Adding the additional 100 kΩresistor would ensure the minimum required input impedance.

For low DC offsets at the output we let Rf1 = 100 kΩ. Solving for Ri (Non-Inverting Amplifier) gives the following:

Ri = Rf1/(AV−1) = 100k/(15 −1) = 7.1 kΩ; use 6.8 kΩ

The bandwidth requirement must be stated as a pole, i.e., the 3 dB frequency. Five times away from a pole gives

0.17 dB down, which is better than the required 0.25 dB. Therefore:

fL= 20 Hz/5 = 4 Hz

fH= 20 kHz × 5 = 100 kHz

At this point, it is a good idea to ensure that the Gain-Bandwidth Product for the part will provide the designed

gain out to the upper 3 dB point of 100 kHz. This is why the minimum GBWP of the LM2876 is important.

GBWP ≥AV× f3 dB = 15 × 100 kHz = 1.5 MHz

GBWP = 2.0 MHz (min) for the LM2876

Solving for the low frequency roll-off capacitor, Ci, we have:

Ci ≥1/(2πRi fL) = 5.9 μF; use 10 μF.

Definition of Terms

Input Offset Voltage: The absolute value of the voltage which must be applied between the input terminals

through two equal resistances to obtain zero output voltage and current.

Input Bias Current: The absolute value of the average of the two input currents with the output voltage and

current at zero.

Input Offset Current: The absolute value of the difference in the two input currents with the output voltage and

current at zero.

Input Common-Mode Voltage Range (or Input Voltage Range): The range of voltages on the input terminals

for which the amplifier is operational. Note that the specifications are not ensured over the full common-mode

voltage range unless specifically stated.

Common-Mode Rejection: The ratio of the input common-mode voltage range to the peak-to-peak change in

input offset voltage over this range.

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Power Supply Rejection: The ratio of the change in input offset voltage to the change in power supply voltages

producing it.

Quiescent Supply Current: The current required from the power supply to operate the amplifier with no load

and the output voltage and current at zero.

Slew Rate: The internally limited rate of change in output voltage with a large amplitude step function applied to

the input.

Class B Amplifier: The most common type of audio power amplifier that consists of two output devices each of

which conducts for 180° of the input cycle. The LM2876 is a Quasi −AB type amplifier.

Crossover Distortion: Distortion caused in the output stage of a class B amplifier. It can result from inadequate

bias current providing a dead zone where the output does not respond to the input as the input cycle goes

through its zero crossing point. Also for ICs an inadequate frequency response of the output PNP device can

cause a turn-on delay giving crossover distortion on the negative going transition through zero crossing at the

higher audio frequencies.

THD + N: Total Harmonic Distortion plus Noise refers to the measurement technique in which the fundamental

component is removed by a bandreject (notch) filter and all remaining energy is measured including harmonics

and noise.

Signal-to-Noise Ratio: The ratio of a system's output signal level to the system's output noise level obtained in

the absence of a signal. The output reference signal is either specified or measured at a specified distortion

level.

Continuous Average Output Power: The minimum sine wave continuous average power output in watts (or

dBW) that can be delivered into the rated load, over the rated bandwidth, at the rated maximum total harmonic

distortion.

Music Power: A measurement of the peak output power capability of an amplifier with either a signal duration

sufficiently short that the amplifier power supply does not sag during the measurement, or when high quality

external power supplies are used. This measurement (an IHF standard) assumes that with normal music

program material the amplifier power supplies will sag insignificantly.

Peak Power: Most commonly referred to as the power output capability of an amplifier that can be delivered to

the load; specified by the part's maximum voltage swing.

Headroom: The margin between an actual signal operating level (usually the power rating of the amplifier with

particular supply voltages, a rated load value, and a rated THD + N figure) and the level just before clipping

distortion occurs, expressed in decibels.

Large Signal Voltage Gain: The ratio of the output voltage swing to the differential input voltage required to

drive the output from zero to either swing limit. The output swing limit is the supply voltage less a specified quasi-

saturation voltage. A pulse of short enough duration to minimize thermal effects is used as a measurement

signal.

Output-Current Limit: The output current with a fixed output voltage and a large input overdrive. The limiting

current drops with time once SPiKe protection circuitry is activated.

Output Saturation Threshold (Clipping Point): The output swing limit for a specified input drive beyond that

required for zero output. It is measured with respect to the supply to which the output is swinging.

Output Resistance: The ratio of the change in output voltage to the change in output current with the output

around zero.

Power Dissipation Rating: The power that can be dissipated for a specified time interval without activating the

protection circuitry. For time intervals in excess of 100 ms, dissipation capability is determined by heat sinking of

the IC package rather than by the IC itself.

Thermal Resistance: The peak, junction-temperature rise, per unit of internal power dissipation (units in °C/W),

above the case temperature as measured at the center of the package bottom.

The DC thermal resistance applies when one output transistor is operating continuously. The AC thermal

resistance applies with the output transistors conducting alternately at a high enough frequency that the peak

capability of neither transistor is exceeded.

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Power Bandwidth: The power bandwidth of an audio amplifier is the frequency range over which the amplifier

voltage gain does not fall below 0.707 of the flat band voltage gain specified for a given load and output power.

Power bandwidth also can be measured by the frequencies at which a specified level of distortion is obtained

while the amplifier delivers a power output 3 dB below the rated output. For example, an amplifier rated at 60W

with ≤0.25% THD + N, would make its power bandwidth measured as the difference between the upper and

lower frequencies at which 0.25% distortion was obtained while the amplifier was delivering 30W.

Gain-Bandwidth Product: The Gain-Bandwidth Product is a way of predicting the high-frequency usefulness of

an op amp. The Gain-Bandwidth Product is sometimes called the unity-gain frequency or unity-gain cross

frequency because the open-loop gain characteristic passes through or crosses unity gain at this frequency.

Simply, we have the following relationship: ACL1 × f1= ACL2 × f2

Assuming that at unity-gain (ACL1 = 1 or (0 dB)) fu = fi = GBWP, then we have the following: GBWP = ACL2 × f2

This says that once fu (GBWP) is known for an amplifier, then the open-loop gain can be found at any frequency.

This is also an excellent equation to determine the 3 dB point of a closed-loop gain, assuming that you know the

GBWP of the device. Refer to the diagram on the following page.

Biamplification: The technique of splitting the audio frequency spectrum into two sections and using individual

power amplifiers to drive a separate woofer and tweeter. Crossover frequencies for the amplifiers usually vary

between 500 Hz and 1600 Hz. “Biamping” has the advantages of allowing smaller power amps to produce a

given sound pressure level and reducing distortion effects prodused by overdrive in one part of the frequency

spectrum affecting the other part.

C.C.I.R./A.R.M.:

Literally: International Radio Consultative Committee

Average Responding Meter

This refers to a weighted noise measurement for a Dolby B type noise reduction system. A filter characteristic is

used that gives a closer correlation of the measurement with the subjective annoyance of noise to the ear.

Measurements made with this filter cannot necessarily be related to unweighted noise measurements by some

fixed conversion factor since the answers obtained will depend on the spectrum of the noise source.

S.P.L.: Sound Pressure Level—usually measured with a microphone/meter combination calibrated to a pressure

level of 0.0002 μBars (approximately the threshold hearing level).

S.P.L. = 20 Log 10P/0.0002 dB

where

• P is the R.M.S. sound pressure in microbars. (1 Bar = 1 atmosphere = 14.5 lb/in2= 194 dB S.P.L.).

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REVISION HISTORY

Changes from Revision B (March 2013) to Revision C Page

• Changed layout of National Data Sheet to TI format .......................................................................................................... 24

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PACKAGE OPTION ADDENDUM

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Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status

(1)

Package Type Package

Drawing Pins Package

Qty Eco Plan

(2)

Lead finish/

Ball material

(6)

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

Samples

LM2876TF/NOPB ACTIVE TO-220 NDA 11 20 RoHS-Exempt

& Non-Green SN Level-1-NA-UNLIM -20 to 85 LM2876TF

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance

do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may

reference these types of products as "Pb-Free".

RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.

Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based

flame retardants must also meet the <=1000ppm threshold requirement.

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

(6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two

lines if the finish value exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

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In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

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